High dynamic range time-varying integrated receiver for elimination of off-chip filters

ABSTRACT

A quadrature mixer with an LO input is provided. The quadrature mixer receives a signal having a frequency F LO  and a signal input having a frequency F SIG , and has an output that comprises an output impedance that is high at frequencies of |F LO −F SIG  | and |F LO +F SIG | and low at other. A mixer coupled to the output impedance interacts with the output impedance such that an impedance presented at the signal input is high for signals at F SIG  if F SIG  is a predetermined signal frequency, and low at other frequencies.

RELATED APPLICATIONS

This application claims priority to U.S. Provisional application60/539,702, filed Jan. 28, 2004, which is hereby incorporated byreference for all purposes.

FIELD OF THE INVENTION

The present invention relates to mixers, and more specifically to aquadrature mixer where the impedance presented to a signal input is highfor signals at a desired signal frequency and low at undesiredfrequencies.

BACKGROUND

Design of typical radio-frequency (RF) receivers in cellular mobileterminals are subject to several design constraints. The firstconstraint is limitations on the ability to reliably detect very weaksignals in the desired frequency channel. The second constraint is theability to detect only slightly stronger in-band signals in the presenceof very strong interfering signals. For instance, for the GSM system,the receiver must be able to reliably detect signals with a strength of−108dBm in the absence of interference and a strength of −99dBm while inthe presence of 0dBm interfering signals at an offset of 20 MHz or more.

The most common solution to solving problems caused by very stronginterfering signals has been to make use of very high quality factor (Q)bandpass filters at the input of the receiver. These filters aretypically surface acoustic wave (SAW) filters which pass the receiveband with a typical attenuation of ˜2.5 dB and attenuate out-of-bandsignals (e.g., 10-20 MHz away from the receive band) by about 20 dB.These filters are highly linear and typically result in a reduction ofout-of-band interfering signals to about the same level as in bandinterference (−23dBm).

There are several drawbacks associated with this approach however. Thefirst is that in-band attenuation tends to make it harder to detect weaksignals, creating the need for an even more sensitive receiver after thefilter. More importantly, there is currently no economical way toimplement SAW filters or their equivalents in the same processes as theactive circuits that follow them, which are typically produced usingCMOS or BiCMOS processes and either silicon or silicon germaniumtechnologies. The result is that SAW filters significantly increase thecost and consume equally valuable circuit board area in a typicalhandset. This problem is further exacerbated by the proliferation ofdifferent frequency bands that a mobile handset has to be compatiblewith.

FIG. 1 is a diagram of an exemplary prior art system 100 for providingmultiple band compatibility. Since each band has a different pass-bandand different stop-bands, each band requires a separate SAW filter 102A,and, consequently separate input ports to the separate receiver inputs104A as well as separate outputs from any transmit/receive (T/R) switch106A or similar device.

FIG. 2 is a diagram of a linear-time-varying (LTV) low-pass filter 200in accordance with the prior art. Filter 200 can be built by combiningthree capacitors and two switches, such as might be used in the separateand traditionally unrelated area of switched capacitor filters. Adifferential current of frequency F_(SW) can be driven across ports V₀₊and V⁰⁻. The bandwidth of the resulting filter is equal to(C_(i)/C₀)•F_(SW). Applying a current signal to V_(i) at a frequencyF_(SW)+dF, where dF is an offset frequency, results in a differentialoutput voltage (V₀₊-V⁰⁻) with a frequency dF and a filtered amplitudewith bandwidth (Ci/Co)•F_(SW). The input voltage V_(i) is partiallyfiltered by an LTV band-pass filter centered on F_(SW) with a bandwidthof 2•(C_(i)/C₀)•F_(SW).

SUMMARY OF THE INVENTION

In accordance with the present invention, a quadrature mixer is providedthat overcomes known problems with quadrature mixers.

In particular, a quadrature mixer is provided that provides a highimpedance at the signal input if the signal is a predetermined signalfrequency, and a low impedance at other frequencies.

In accordance with an exemplary embodiment of the present invention, aquadrature mixer with a local oscillator (LO) input is provided. Thequadrature mixer receives a signal having a frequency F_(LO) and asignal input having a frequency F_(SIG), and has an output thatcomprises an output impedance that is high at frequencies of|F_(LO)−F_(SIG)| and |F_(LO)+F_(SIG)| and low at other. A mixer coupledto the output impedance interacts with the output impedance such that animpedance presented at the signal input is high for signals at F_(SIG)if F_(SIG) is a predetermined signal frequency, and low at otherfrequencies.

The present invention provides many important technical advantages. Oneimportant technical advantage of the present invention is its ability toincrease the wide-band linearity of an otherwise standard radioreceiver. This in turn permits implementations without external high-Qfilters, significantly reducing cost.

Those skilled in the art will further appreciate the advantages andsuperior features of the invention together with other important aspectsthereof on reading the detailed description that follows in conjunctionwith the drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a diagram of an exemplary prior art system for providingmultiple band compatibility;

FIG. 2 is a diagram of a linear-time-varying (LTV) low-pass filter inaccordance with the prior art;

FIG. 3A is a diagram of an exemplary system for providing quadratureband GSM compatibility using the invention described herein;

FIG. 3B is a diagram of a system that uses frequency selectivity inaccordance with an exemplary embodiment of the present invention;

FIG. 4 is a diagram of a system for high-performance applications thatuse an additional input LC resonator, in accordance with an exemplaryembodiment of the present invention;

FIGS. 5A through 5C are charts 500, 502 and 504 demonstrating howlinearity can be degraded as a consequence of the time-varying inputresistance of a MOSFET-based passive mixer;

FIG. 6 is a chart of the phase dependent resistance that results from arepresentative I/Q mixer with inputs connected together, in accordancewith an exemplary embodiment of the present invention;

FIG. 7 is a diagram of a system in accordance with an exemplaryembodiment of the present invention;

FIG. 8 is a diagram of a system in accordance with an exemplaryembodiment of the present invention;

FIG. 9 is a diagram of switch resistance as a consequence of inputsignal;

FIG. 10 is a diagram of complementary CMOS switches for improving mixerlinearity in accordance with an exemplary embodiment of the presentinvention;

FIGS. 11A through 11C are charts showing circuit parameters inaccordance with an exemplary embodiment of the present invention;

FIG. 12 is a diagram of an exemplary drive structure in accordance withan exemplary embodiment of the present invention; and

FIG. 13 is a diagram of a circuit for performing a balun function whilesimultaneously performing impedance matching to the LNA input inaccordance with an exemplary embodiment of the present invention.

DETAILED DESCRIPTION OF PREFERRED EMBODIMENTS

In the description that follows, like parts are marked throughout thespecification and drawings with the same reference numerals,respectively. The drawing figures might not be to scale, and certaincomponents can be shown in generalized or schematic form and identifiedby commercial designations in the interest of clarity and conciseness.

In one exemplary embodiment, the present invention permits detection ofweak in-band signals in the presence of very strong out-of-bandinterference on a CMOS integrated circuit without the need for externalhigh-Q filtering, such as may conventionally be achieved using SAWfilters. These circuits can also be combined with a suitable LO topermit multiband operation, such as full quad-band GSM/DCS/PCSoperation. In addition, the relatively wide bandwidth of the inputcombined with the elimination of the conventional requirement forexternal filters allows a reduction of the number of inputs required forthe receiver in addition to a similar reduction in the required outputsfrom a T/R switch. For example, in a GSM/DCS/PCS handset, two taps couldbe used instead of the conventional requirement for three.

In another exemplary embodiment, the front-end of the receiver can beimplemented without a SAW filter or other similar components byconnecting a low noise amplifier (LNA) and a passive mixer. As usedherein, a connection can include a direct connection, a connectionthrough one or more intervening components, or other suitableconnections. This configuration down-converts weak RF signals in thepresence of strong out-of-band blocking signals. The LNA can include ahighly linear transconductor or other suitable components that allow thecurrent signal delivered to the LNA load to be provided with acceptablylow distortion from any transconductance nonlinearities. The output ofthe LNA is connected to the input of the mixer so as to use the mixer'stime varying properties to down convert the received signal to alow-intermediate frequency (IF) or baseband signal. Additionally, theinput of the mixer acts as part of the LNA's load network. Thisconnection allows the input impedance of the mixer, which acts abandpass filter due to the time varying properties of the mixer, tofilter the output voltage of the LNA. This filtering action reduces theamplitude of the unwanted interference signals as well as increases thelinearity of the LNA by reducing the voltage levels caused byinterference signals which might otherwise drive the LNA into anonlinear mode of operation, which can also result in intermodulationand/or gain reduction of the desired receive signal.

In another exemplary embodiment, the LNA can be a pseudo-differential,heavily degenerated common source amplifier with asingle-ended-to-differential converting matching network. Alternatively,a single-ended LNA, a fully differential LNA, or other suitable LNAcircuits may be used. The LNA load network can include a parallelinductor/capacitor (LC) network connected in parallel with the passivemixer input. An RF choke in parallel with the mixer input, a transformerconnected between the LNA and the mixer input, or other suitable loadnetworks can also or alternatively be used.

The passive mixer can be any suitable mixer that provides apredetermined bandpass response to a signal provided to its input port.In one implementation, the mixer can include two double balanced passivemixers driven by quadrature LO signals. A suitable technique is used toensure high linearity and reliability in the mixer switches. Forexample, the mixer output load can include one or several largecapacitors which act to set the effective bandwidth of the mixer. Thisbandwidth affects not only the output low-pass bandwidth of the mixer,but sets the bandwidth of a band-pass-like response that appears at theinput of the mixer, centered close to the LO or reference frequency. Theresulting band-pass characteristic attenuates interfering signals thatare separated by a large frequency difference so as to not overload themixer and subsequent circuitry, as well as to help ensure that suchinterfering signals do not overload the output of the LNA.

The linearity, gain and noise figures can also be further improved bythe use of LC tanks placed in series with the output(s), such as wherethe LC tanks are tuned to approximately twice the LO frequency.

FIG. 3A is a diagram of an exemplary system 300A for providing quad-bandGSM compatibility without filters by using a mixer 105B described here.

FIG. 3B is a diagram of a system 300B that uses frequency selectivity inaccordance with an exemplary embodiment of the present invention. System300B avoids the need for low-Q LC resonator-based filters, such as thosethat are typically used to achieve an LNA input match, and thisfrequency selectivity is instead postponed until after the LNA stage. Inorder to accomplish this, the LNA input should be more linear than aconventional receiver, e.g. about 20 dB more linear than in aconventional GSM receiver. In one exemplary embodiment, the linearitycan be improved partly by increasing the noise figure, as the noisefigure is no longer degraded by the in-band attenuation of the input SAWfilter. The output linearity of the LNA, however, can present a muchgreater challenge to signal processing as any already strong blockingsignal from the input would have been amplified by typically between 10dB and 20 dB. For example, a 0dBm signal on the input would result inapproximately +15dBm at the output, or about a 4 volt peak swing (for atypical on-chip impedance of 200Ω), which is a prohibitively large swingfor a linear RF integrated circuit. This problem can be overcome bypresenting the LNA output with a high-Q band-select output impedance,centered on the wanted frequency and with a sufficiently narrowbandwidth so that high-power interferes outside of this bandwidth willbe attenuated to voltage levels where the LNA output is sufficientlylinear.

Received signals pass first through impedance match 314, stepping upvoltage swing while decreasing current swing to better drive the LNA316. This LNA amplifies the signal from the impedance match and drivesit, in current mode onto a load consisting of an LC tank made up ofcapacitor 318 and inductor 320 in parallel with two passive mixers 302Aand 302B. These passive mixers are switched by local oscillator signals;mixer 302A is switched by signal from buffer 310, at 0 degrees phase,and mixer 302B is switched by signal from buffer 312, at 90 degreesphase. Each mixer 302A and 302B is loaded by a large capacitor (308A and308B respectively) and then drives a baseband chain of additionalprocessing (304 and 306 respectively).

The high-Q impedance is provided by the time varying behavior of thepassive mixers 302A and 302B that follows the LNA. As discussed below,passive switching mixers 302A and 302B can act to convolve the frequencyresponse of its output impedance with its LO frequency or switchingfrequency to provide a tuned high-Q filter centered on the switchingfrequency. The resulting time varying input impedance of the mixers 302Aand 302B, in combination with any other loading circuitry sets theoutput load of the LNA and thus sets its gain. Since this impedancetakes the form of a narrow-band peak, signals out of this band will tendto be attenuated at the LNA output relative to in-band signals resultingin reduced linearity requirements for the LNA output and subsequentcircuitry.

Concurrent with this filtering action, the mixer acts to down-convertits input to low-pass filtered baseband or low-IF signal(s). In oneexemplary embodiment, the mixer can be configured to produce in-phaseoutput 304 and quadrature output 306 signals. In another embodiment,high-voltage-swing LO signals are used to drive the mixer so that theswitches operate in a highly linear mode. This high voltage LOimplementation, in turn, requires specialized high-swing drivers andmixer biasing techniques to guarantee reliability under high swing.

One property of passive mixers such as switching mixers 302A and 302B isthat they are bi-directional and so tend to generate a time-varyinginteraction between their input and output impedances. Similar to theswitched-capacitor effect described above, this means that largecapacitors 308A and 308B placed on the output of passive mixers 302A and302B interact with the mixer to form an output low-pass filter whichconvolves in the frequency domain with the LO to form an input band-passimpedance response centered on or close to the switching frequency.

FIG. 4 is a diagram of a system 400 for high-performance applicationsthat use an additional input LC resonator, in accordance with anexemplary embodiment of the present invention. System 400 providesadditional filtering as well as keeping the capacitance connected to theinput from detuning the response. As in the case for the simplecapacitive loading, this circuit provides a high-Q input band-passimpedance.

The benefit, as discussed in previous sections, is that the band-passform of the input impedance can be used to attenuate out-of-band currentmode signals before they are translated into voltages, therebyincreasing the out-of-band signal strength which can be applied at theinput of the LNA before driving the circuitry into nonlinearity.Unfortunately, simple switched capacitive structures, even indifferential implementations such as shown in FIG. 4, are poorly suitedto high-performance applications such as cellular handsets. Theseproblems primarily involve mixer linearity and the generation ofquadrature outputs as is typically required.

Inductor 402 and capacitor 404 form a parallel resonant tank, with apeak impedance at or near the desired receive frequency. This tank is inparallel with the passive mixer, made up of switches 406, 408, 410 and412. At any given time either switches 408 and 412 are closed, orswitches 406 and 410 are closed, connecting the tank across Co inalternating polarities every half cycle of V_(SW).

FIGS. 5A through 5C are charts 500, 502 and 504 demonstrating howlinearity can be degraded as a consequence of the time-varying inputresistance of a MOSFET-based passive mixer. The consequence is that theeffective average impedance seen by an input signal near the LOfrequency will depend on the phase relationship between LO and RF, asdepicted in FIG. 5B and FIG. 5C. For an RF frequency F_(RF) which isdifferent from the LO frequency F_(LO) by an offset dF such thatF_(RF)=F_(LO)+dF, this results in amplitude modulation on the RF signalat frequency dF, potentially leading to second-order nonlinearityeffects and effectively reducing the desired blocker attenuation.Additionally, nonlinearity can occur when using FET switching devices inthe mixer if a significant drain-source voltage V_(ds) appears acrossthe switching devices. Due to a number of physical effects in FETtransistors, such as channel pinch off near the drain region andvelocity saturation for short channel devices, the resistance of a FETswitch is typically a function of the channel voltage, resulting in anonlinearity between the switch voltage and current. This nonlinearitycan be reduced by keeping the V_(ds) of the device as low as possible,since the resistance becomes more linear as the V_(ds) amplitudeapproaches zero. This problem can be partially solved by making thenominal R_(ch) low, but additional methods to reduce V_(ds) could alsoor alternatively be used.

FIG. 6 is a chart 600 of the phase dependent resistance that resultsfrom a representative I/Q mixer with inputs connected together, inaccordance with an exemplary embodiment of the present invention. In thecase where in-phase (I) and quadrature (Q) signals are required, themixer section can consist of two mixers driven by in-phase andquadrature LO signals, respectively. In conventional implementations,some isolation between the two mixing cores is typically desired toavoid interaction between the mixing actions of the two mixers. In thiscase, however, several benefits are obtained by tying the RF inputs ofthe I and Q mixers together. The first benefit is simplicity, whichresults in lower power consumption since all active input circuits(e.g., the LNA) can be shared. The second effect is that since the I andQ mixers are 90 degrees out of phase, the second harmonic component ofthe time varying input resistance of the two mixers tend to cancel oneanother, which removes the phase-dependant resistance described above,and thereby reduces inadvertent sideband generation.

FIG. 7 is a diagram of a system 700 in accordance with an exemplaryembodiment of the present invention. For the first quarter cycle of theLO, switches 702 and 708 are on. Then, for the second quarter of thecycle of the LO, switches 702 and 704 are on. For the third quarter ofthe cycle for the LO, switches 704 and 708 are, and for the last quarterof a cycle of the LO, switches 706 and 708 are on. One drawback of thisconfiguration is that at any given moment, the outputs of the I and Qmixers are shorted together. Since the exact configuration of what isshorted changes every quarter of an LO cycle, charge sharing between theoutput capacitors can generate a rapid leakage path. For example, in thefirst quarter cycle, CI+ is shorted to CQ− and CI− to CQ+, so each pairhas equal voltage, and in the second quarter cycle CI+ is shorted to CQ+and CI− to CQ−. In order for this change to occur, a charge of (VI/2)•CQmust be transferred from CI+ to CI−, which would require a rapiddischarge of the output capacitors, a severe reduction in low frequencygain and a massive broadening of the effective bandwidth, all of whichare unacceptable. Hence, in the process of overcoming the problems ofphase-dependant impedance and quadrature down-conversion, a new problemthat has not been previously identified and which must be addressed isintroduced: leakage between I and Q.

An alternative way of describing the leakage problem is to notice thatthis leakage occurs at the frequency 2•F_(LO). One can view theimpedance seen at the input of the mixer Zi(ω_(RF)) as the parallelcombination Z_(o)(ω_(LO)−ω_(RF)) ||Z_(o)(ω_(LO)+ω_(RF)), where Z_(o)(ω)is the impedance seen by the output of the mixer at frequency ω. Placinglarge capacitors on the outputs of the mixer means thatZ_(o)(ω_(LO)−ω_(RF))→∞ as ω_(RF)→ω_(LO), but results in a very lowimpedance for Z_(o)(ω_(LO)+ω_(RF)). The low impedance, high frequencyterm tends to shunt the high-impedance low frequency term, which reducesgain and increases bandwidth. This effect does not appear for typicalsingle phase (non-quadrature) cases because the second-harmonic shuntingeffect only acts when the instantaneous phase between RF and the LOapproaches 90 degrees, and so acts primarily when the down-convertedsignal is passing through zero anyway.

FIG. 8 is a diagram of a system 800 in accordance with an exemplaryembodiment of the present invention. Operation of switches 802, 804, 806and 808 occurs as previously described in regards to switches 702through 708 of FIG. 7. By adding resonant parallel LC tanks 810, 812,814 and 816 in series with the large output capacitors CI and CQ andtuning them to approximately 2•F_(LO), the impedance can be made to gohigh near both F_(LO)−F_(RF) and F_(LO)+F_(RF).

By using a value of L•Q•(ω_(LO)+ω_(RF)) for small values of|ω_(LO)−ω_(RF)| and 1/(C_(L)•|ω_(LO)−ω_(RF)|) for larger values, theresonant tanks can be used to recover the frequency selectivity of themixer while permitting quadrature down-conversion. A second benefit ofadding these tanks is that they permit the drains of the switches thatare on to track their sources in an RF sense (while still tracking theoutput capacitance in a low frequency sense), such that the result is areduction in signal-dependant V_(ds), and thus increased linearity.

FIGS. 11A through 11C are charts 1100A through 1100C showing circuitparameters in accordance with the exemplary embodiment of the presentinvention as shown in FIG. 8. Chart 1100A shows an exemplary diagram ofrelevant impedances of a mixer where Zbb=1/sCo, Zin, Z_(2LO) are tunedRLCs.

FIG. 11B shows an exemplary chart 1100B of output impedance as afunction of a combination of Zbb, Zin and Rch. These are the impedancesof 1100A but with Zin translated by F_(LO) to be centered on baseband.This figure demonstrates the effect of Zin on effective Zout.

FIG. 11C shows an exemplary chart 1100C of input impedance based on acombination of Zin, Rch , frequency-shifted Zbb and Z_(2LO). This chartdemonstrates how interactions between Zin and the translated Z_(2LO) setthe maximum effective Zin. Furthermore it demonstrates how theseimpedances interact with the translated Z_(bb) sets intput bandwidth.Finally this chart shows how R_(ch) limits wideband attenuation of theeffective Zin.

In implementations using FET devices such as MOSFETs, the point when aswitch turns on as well as its conductance during the on-state bothdepend upon not just the LO-driven gate voltage, but also the inputsource voltage and output drain voltage. One consequence is that stronginput signals can modulate the switching point of a transistor, as shownin the effective mixing waveform of FIG. 9. A second effect is thatstrong input signals can modulate gds and thus Ron of the switches,which in turn changes the degree of blocker signal attenuation. Both ofthese effects result in nonlinearity in the mixer, degradingperformance.

FIG. 9 is a graph 900 showing the effect of source (input) voltage onresistance of a given switch. Since peak resistance 901 relates to theattenuation of out-of band signals, changes in resistance due to largesignals imply second-order nonlinearity effecting large out of bandsignals, which is highly undesirable. Furthermore, it can be seen thatchanges in Vs change the turn-on-time of the switch 902. Such modulationof timing affects the properties of the mixer in ways that affect bothwanted and unwanted signals.

FIG. 10 is a diagram of complementary CMOS switches 1000 for improvingmixer linearity in accordance with an exemplary embodiment of thepresent invention. The configuration of complementary CMOS switches 1000allows the PMOS switches 1002A through 1002D to respond to changes insource voltage Vs in the complementary way to the NMOS switches 1004Athrough 1004D, such that to first order, changes in transistorconductance due to large signals on the RF input will cancel and lead toa conductance which tends to be relatively independent of source anddrain voltages, improving linearity. Similarly, changes in the effectivetransition points of the individual switching transistors will haveopposite polarity and so their effects tend to cancel to first order.Although using complimentary switches increases the parasiticcapacitance of the mixer on all of its ports, it does not necessarilyrequire extra circuitry to drive. In a differential mixer, complimentaryLO signals are typically required even without complementary switches todrive the two differential phases, and so the polarity of the NMOS andPMOS LO drives can simply be reversed. Typically, PMOS transistorsshould be scaled relative to NMOS such that they have equalconductances, i.e. equal values of μ•C_(ox)•W/L.

An additional problem to be addressed is that the series resistance ofthe mixer switches, when on, must be relatively low, such as to avoidexcessive noise generation in those switches and other problems. Theseries resistance of the switches also sets the limit on the amountout-of-band attenuation which is possible to achieve at the input of themixer, where the inband-to-out-of-band attenuation ratio isapproximately equal to the ratio between the impedance seen by the inputof the mixer to the series resistance of the mixer switches, as shown inFIG. 11C. Using short channel length transistors, with as much totalgate width as possible and with the largest possible gate drive voltagewhen on, can avoid these problems.

High gate drive swing is desirable to minimize the series resistance ofthe mixer switches. In addition, increased swing can improve theeffective linearity of the mixer by reducing the ratio between the RFand LO voltages, V_(RF)/V_(LO), which is approximately equal to theratio between the difference in source voltage induced by the rd and thedc gate-source voltage imposed by the LO, dV_(s)/V_(gs). This ratio isroughly proportional to the effect that RF signals have on theconductance of the mixer switches, and so directly relates to linearity.Likewise, since higher swing implies faster transitions, higher swingimplies that changes in Vs have less effect on transition times, alsorelating to linearity. Ideally it would be possible to drive V_(gs) ashigh as is possible, which for CMOS devices is close to the oxidebreak-down voltage V_(box). Since the LO drive signal will typically beapproximately sinusoidal and so have equal maximum and minimumdeviations from its bias point, any bias voltage difference between thegates (LO inputs) and other ports (the source, drain, and bulk) will addto this voltage difference and so decrease the maximum possible swingpossible without breaking the transistor down. Thus it is usuallybeneficial to provide that the gate, drain, source and bulk take on thesame voltage. In processes that allow both N-type and P-type transistorsto be dc isolated from the bulk, such as triple-well orsilicon-on-insulator (SOI), this can be achieved by choosing a biasvoltage (typically based upon the required output level) and tying thebulks directly to it, while tying the drains (baseband outputs) andgates to that voltage through large resistances, and then AC coupling inthe RF and LO signals.

FIG. 12 is a diagram of an exemplary drive structure 1200 in accordancewith an exemplary embodiment of the present invention.

One difficulty in achieving large LO voltage swing (e.g., LOswing=2•V_(box)) is that the LO drive circuit must produce this signal.A purely active approach would face the problem that any output devicewill inherently see this large voltage swing (e.g., 2•V_(box)), and willbe prone to break down. If the mixer is implemented with FET deviceswith the gates driven by the LO, the LO input impedance is largelycapacitive.

The LO buffer consists of a pair of CMOS inverters 1202A and 1202B,driven differentially and each consisting of NMOS transistors 1206 and1210 and PMOS transistors 1208 1212. The outputs of the inverters 1202Aand 1202B drive the mixer 1216 differentially through two inductors1204A and 1204B respectively. These inductors resonate with any inputcapacitance to generate increased voltage swing. The amplitude of thisswing is set by the available charge for each half-cycle of the buffer.Charge is stored on capacitor 1214, which is continuously charged byIbias and discharged alternately by inverters 1202A and 1202B.

Complementary push-pull drivers 1202A and 1202B can be used to achievereasonable efficiency, while an inductor 1204A and 1204B placed inseries with the output with a value chosen to resonate with the LO inputcapacitance of the mixer permits a voltage step-up by a factor ofapproximately Q, where Q is the quality factor of the LCR resonatorconsisting of the inductor and the mixer input capacitance. By choosinga Q>2 the swing at the output of the driver can be kept less than thebreakdown voltage while driving the desired peak-to-peak swing (e.g.,approximately 2•V_(box)). Optionally, an additional technique can beused to make the output swing independent of the resonant Q. By currentlimiting the push-pull stage, one can guarantee a peak-to peak swing ofI_(bias)/(C_(in)•F_(LO)) regardless of inductor Q (permitting the use ofa higher Q inductor without worrying about overdriving the mixer. Notethat Q must be at least 2 to allow a peak-to-peak mixer drive of V_(box)without presenting the driver with more than V_(box).

FIG. 13 is a diagram of a circuit 1300 for performing a balun functionwhile simultaneously performing impedance matching to the LNA input inaccordance with an exemplary embodiment of the present invention.

Circuit 1300 includes a matching circuit 1302 and a transaconductance1304. The differential LNA and matching network of FIG. 13 works asfollows. A pair of signals containing both differential and common-modecomponents is AC-coupled into the matching network through capacitors1306 and 1308. Parallel resonance set up by the combined inductance of1312 and 1314 and capacitances 1310, 1306 and 1308 act to increasevoltage swing in the differential mode. a series resonance set up byinductors 1312 and 1314 and capacitor 1316 acts to suppress common modevoltage swing. This differential voltage signal is presented to thegates of transistors 1322 and 1324, which act as a psuedo-differentialamplifier, and which are linearized by the degeneration presented byinductors 1318 and 1320. The current-mode signals for transistors 1322and 1324 are then buffered by cascode transistors 1326 and 1328.

Achievable LNAs will typically be relatively broad-band, compared topossible in-band interfering frequency separation and so must besufficiently linear to handle the maximum blocking signal strengthwithout unacceptable degradation in amplification of weak wantedsignals. This condition imposes two requirements for the LNA design. Thefirst is that the LNA should have a very high input compression point;the second is that the LNA should have a high degree of rejection foreven-order nonlinearities, as these tend to cause misoperation in low-IFand direct conversion receivers. When using a common-source styleamplifier, high input compression implies a large amount of sourcedegeneration. For reasons of noise and input matching, a relativelylarge inductance in series with the source combined with a relativelyhigh bias current is preferable. Specifically, I_(bias)•L•ω>V_(swing),where I_(bias) is the bias current, L is the source inductor, and ω isthe operating frequency. For suppression of even-order nonlinearities,one approach is to use a differential LNA. This helps reducenonlinearity both in the LNA itself and, if the following mixer is alsodifferential, the even order linearity of the mixer which it drives. Ifthe input signal must be single-ended, as is commonly the case, somesort of single-ended to differential conversion is needed.

Although exemplary embodiments of a system and method of the presentinvention have been described in detail herein, those skilled in the artwill also recognize that various substitutions and modifications can bemade to the systems and methods without departing from the scope andspirit of the appended claims.

1. A receiver circuit comprising: a quadrature passive mixer having aninput and an output; one or more output impedances coupled to the outputof the quadrature passive mixer wherein an input impedance of thequadrature passive mixer provides a band-pass response; a low noiseamplifier (LNA) having an input and an output coupled to the quadraturepassive mixer, the LNA configured to provide substantially lineartransconductance over a predetermined input range; and wherein the LNAand the quadrature passive mixer are configured so that a band-passmixer input impedance response is presented to the output of the LNA soas to substantially reduce the LNA voltage gain for unwanted signalspresented to the LNA input.
 2. The receiver of claim 1, wherein the LNAis a common source amplifier and includes source degeneration.
 3. Thereceiver of claim 1, wherein the LNA is a differential amplifier.
 4. Thereceiver of claim 1 wherein: a frequency of operation corresponds to achannel in a wireless voice communications band; the lineartransconductance of the LNA input reduces degradation of an in-bandsignal in the presence of an out-of band signal; and the band-passaction of the mixer attenuates the out-of band signals so as to preventthe out-of-band signals from degrading the in-band signal at the outputof the LNA.
 5. The receiver of claim 4 wherein the receiver is selectedfrom one or more of the group comprising a low intermediate frequencyreceiver and a direct conversion receiver.
 6. The receiver of claim 4wherein: the desired frequency corresponds to a channel in a GSM band.the LNA input is sufficiently linear so as to not significantly degradesensitivity-level wanted signals in the presence of out-of band signals;the band-pass action of the mixer attenuates out-of band signals toprevent out-of-band signals from degrading wanted signals at the outputof the LNA.
 7. The mixer of claim 2, further comprising: a plurality ofNFETs and PFETs; and wherein the NFETs and PFETs are driven withsubstantially differential signals.
 8. The quadrature mixer of claim 2,further comprising: a plurality of FET switches; and wherein two or moreof the FET switches are biased to approximately the same voltage.
 9. Thequadrature mixer of claim 2, wherein: the LO inputs of one or moreswitches of the mixer are driven by buffers through series-resonant LCstructures; the LC structures are tuned to the LO frequency; and thepeak-to-peak swing at the switches is enhanced beyond the breakdownvoltage of the buffer.
 10. A quadrature mixer comprising: a localoscillator (LO) input receiving a signal having a frequency F_(LO); asignal input receiving a signal having a frequency F_(SIG); an outputimpedance that is high at frequencies of |F_(LO)−F_(SIG)| and|F_(LO)+F_(SIG)| and low at other frequencies; and a mixer coupled tothe output impedance, the LO and the signal input, wherein an impedancepresented at the signal input is high for signals at F_(SIG) if F_(SIG)is a predetermined signal frequency, and low at other frequencies. 11.The quadrature mixer of claim 10, wherein the LNA load comprises themixer input in parallel with a parallel LC resonator.
 12. The quadraturemixer of claim 11, wherein a balun provides single-ended to differentialconversion and impedance matching to the LNA inputs.
 13. A quadraturemixer comprising: an input; an in-phase output; a quadrature output; alocal oscillator (LO) generating a frequency and driving the quadraturemixer near a desired input signal frequency; two load networks, each ofwhich presents a low pass impedance response near zero frequency and ahigh impedance response at twice the LO frequency, wherein each of thein-phase output and the quadrature phase outputs are connected to one ofthe load networks; and wherein each of the load networks interacts withtime varying properties of the quadrature mixer to provide a band-passresponse at the input.
 14. The quadrature mixer of claim 13 wherein thein-phase output and the quadrature-phase output each comprises: a mixerhaving an input connected to the quadrature mixer input and two outputs;two parallel LC resonators, each with a resonant frequency of twice thefrequency of the local oscillator, and each having a first terminalconnected to one of outputs of the mixer, and a second terminal; twocapacitors, each having a first terminal and a second terminal, whereinthe first terminal and second terminal of each capacitor is connected tothe second terminals of each parallel RC resonator, wherein eachcapacitor modifies a bandwidth of the mixer's band-pass response.
 15. Adifferential quadrature mixer comprising: an input; an in-phasedifferential output; a quadrature differential output; a localoscillator (LO) generating a frequency and driving the differentialquadrature mixer near a desired input signal frequency; each of thedifferential outputs is connected to a load network which presents a lowpass impedance response near zero frequency and a high impedanceresponse at twice the LO frequency; and each of the load networksinteracts with time varying properties of the differential quadraturemixer to provide a band-pass response at the input.
 16. The differentialquadrature mixer of claim 15 wherein the load network for eachdifferential output comprises: two parallel LC resonators with aresonant frequency of twice the frequency of the LO, each having a firstterminal connected to one of the differential outputs and a secondterminal; a capacitor having a first terminal and a second terminal,each terminal connected to one of the second terminals of one of the twoparallel LC resonators; and wherein the capacitors interact with thetime-varying properties of the differential quadrature mixer todetermine a bandwidth of a band-pass input impedance of the differentialquadrature mixer.
 17. The differential quadrature mixer of claim 15wherein a signal present at the terminals of the capacitors provides anoutput signal used by successive circuits.
 18. The mixer of claim 15further comprising: a plurality of NFET and PFET switches; and whereinthe NFETs and PFETs are driven with substantially differential signals.19. The quadrature mixer of claim 15 further comprising: two or moreFETs; and wherein the two or more FETs are biased to approximately thesame voltage.
 20. The mixers of claim 15, further comprising: aplurality of switches, each having LO inputs that are driven by buffersthrough series-resonant LC structures; and wherein the LC structures aretuned to the LO frequency, and the peak-to-peak swing at the switches isenhanced beyond the breakdown voltage of the buffer.
 21. A passive mixerdriven at a local oscillator (LO) frequency, comprising: an input; afirst unfiltered output; a second unfiltered output; a first filteredoutput; a second filtered output; the first unfiltered output isconnected to the first filtered output through a first parallel resonantLC tank; the second unfiltered output is connected to the secondfiltered output through a second parallel resonant LC tank; the firstfiltered output is connected to the second filtered output through acapacitor; the first parallel resonant LC tank and the second parallelresonant LC tank are each tuned to provide high impedance at twice theLO frequency; and the first parallel resonant LC tank, the secondparallel resonant LC tank, and the capacitor interact to provide aband-pass input impedance at the input.